SBOS511B April 2015 – December 2015
The INA250 features a 2-mΩ, precision, current-sensing resistor and a 36-V common-mode, zero-drift topology, precision, current-sensing amplifier integrated into a single package. High precision measurements are enabled through the matching of the shunt resistor value and the current-sensing amplifier gain providing a highly-accurate, system-calibrated solution. Multiple gain versions are available to allow for the optimization of the desired full-scale output voltage based on the target current range expected in the application.
The INA250 features a precise, low-drift, current-sensing resistor to allow for precision measurements over the entire specified temperature range of –40°C to 125°C. The integrated current-sensing resistor ensures measurement stability over temperature as well as improving layout and board constraint difficulties common in high precision measurements.
The onboard current-sensing resistor is designed as a 4-wire (or Kelvin) connected resistor that enables accurate measurements through a force-sense connection. Connecting the amplifier inputs pins (VIN– and VIN+) to the sense pins of the shunt resistor (SH– and SH+) eliminates many of the parasitic impedances commonly found in typical very-low sensing-resistor level measurements. Although the sense connection of the current-sensing resistor can be accessed via the SH+ and SH– pins, this resistor is not intended to be used as a stand-alone component. The INA250 is system-calibrated to ensure that the current-sensing resistor and current-sensing amplifier are both precisely matched to one another. Use of the shunt resistor without the onboard amplifier results in a current-sensing resistor tolerance of approximately 5%. To achieve the optimized system gain specification, the onboard sensing resistor must be used with the internal current-sensing amplifier.
The INA250 has approximately 4.5 mΩ of package resistance. 2 mΩ of this total package resistance is a precisely-controlled resistance from the Kelvin-connected current-sensing resistor used by the amplifier. The power dissipation requirements of the system and package are based on the total 4.5-mΩ package resistance between the IN+ and IN– pins. The heat dissipated across the package when current flows through the device ultimately determines the maximum current that can be safely handled by the package. The current consumption of the silicon is relatively low, leaving the total package resistance carrying the high load current as the primary contributor to the total power dissipation of the package. The maximum safe-operating current level is set to ensure that the heat dissipated across the package is limited so that no damage to the resistor or the package itself occurs or that the internal junction temperature of the silicon does not exceed a 150°C limit.
External factors (such as ambient temperature, external air flow, and PCB layout) can contribute to how effectively the heat developed as a result of the current flowing through the total package resistance can be removed from within the device. Under the conditions of no air flow, a maximum ambient temperature of 85°C, and 1-oz. copper input power planes, the INA250 can accommodate continuous current levels up to 15 A. As shown in Figure 30, the current handling capability is derated at temperatures above the 85°C level with safe operation up to 10 A at a 125°C ambient temperature. With air flow and larger 2-oz. copper input power planes, the INA250 can safely accommodate continuous current levels up to 15 A over the entire –40°C to 125°C temperature range.
The INA250 features a physical shunt resistance that is able to withstand current levels higher than the continuous handling limit of 15 A without sustaining damage to the current-sensing resistor or the current-sensing amplifier if the excursions are very brief. Figure 31 shows the short-circuit duration curve for the INA250.
System calibration is common for many industrial applications to eliminate initial component and system-level errors that can be present. A system-level calibration can reduce the initial accuracy requirement for many of the individual components because the errors associated with these components are effectively eliminated through the calibration procedure. Performing this calibration can enable precision measurements at the temperature in which the system is calibrated, but as the system temperature changes as a result of external ambient changes or due to self heating, measurement errors are reintroduced. Without accurate temperature compensation used in addition to the initial adjustment, the calibration procedure is not effective in accounting for these temperature-induced changes. One of the primary benefits of the very low temperature coefficient of the INA250 (including both the integrated current-sensing resistor and current-sensing amplifier) is ensuring that the device measurement remains highly accurate, even when the temperature changes throughout the specified temperature range of the device.
For the integrated current-sensing resistor, the drift performance is shown in Figure 32. Although several temperature ranges are specified in the Electrical Characteristics table, applications operating in ranges other than those described can use Figure 32 to determine how much variance in the shunt resistor value can be expected. As with any resistive element, the tolerance of the component varies when exposed to different temperature conditions. For the current-sensing resistor integrated in the INA250, the resistor does vary slightly more when operated in temperatures ranging from –40°C to 0°C than when operated from 0°C to 125°C. However, even in the –40°C to 0°C temperature range, the drift is still quite low at 25 ppm/°C.
An additional aspect to consider is that when current flows through the current-sensing resistor, power is dissipated across this component. This dissipated power results in an increase in the internal temperature of the package, including the integrated sensing resistor. This resistor self-heating effect results in an increase of the resistor temperature helping to move the component out of the colder, wider drift temperature region.
The INA250 current-sense amplifier can be configured to measure both unidirectional and bidirectional currents through the reference voltage level applied to the reference pin, REF. The reference voltage connected to REF sets the output level that corresponds with a zero input current condition. For unidirectional operation, tie the REF pin to ground so that when the current increases, the output signal also increases upwards from this reference voltage (or ground in this case). For bidirectional currents, an external voltage source can be used as the reference voltage connected to the REF pin to bias up the output. Set the reference voltage to enable sufficient range above and below this level based on the expected current range to be measured. Positive currents result in an output signal that increases from the zero-current output level set by the reference voltage whereas negative currents result in an output signal that decreases.
For both unidirectional and bidirectional applications, the amplifier transfer function is shown in Equation 1:
As with any difference amplifier, the INA250 common-mode rejection ratio is affected by any impedance present at the REF input. This concern is not a problem when the REF pin is connected directly to a reference or power supply. When using resistive dividers from a power supply or a reference voltage, buffer the REF pin with an op amp.
An obvious and straightforward location for filtering is at the device output; however, this location negates the advantage of the low output impedance of the output stage buffer. The input then represents the best location for implementing external filtering. Figure 33 shows the typical implementation of the input filter for the device.
The addition of external series resistance at the input pins to the amplifier, however, creates an additional error in the measurement. Keep the value of these series resistors to 10 Ω or less, if possible, to reduce the affect to accuracy. The internal bias network illustrated in Figure 33 present at the input pins creates a mismatch in input bias currents when a differential voltage is applied between the input pins, as shown in Figure 34.
If additional external series filter resistors are added to the circuit, the mismatch in bias currents results in a mismatch of voltage drops across the filter resistors. This mismatch creates a differential error voltage that subtracts from the voltage developed across the Kelvin connection of the shunt resistor, thus reducing the voltage that reaches the amplifier input terminals. Without the additional series resistance, the mismatch in input bias currents has little effect on device operation as a result of the low input bias current of the amplifier and the typically low impedance of the traces between the shunt and amplifier input pins. The amount of error these external filter resistors add to the measurement can be calculated using Equation 3, where the gain error factor is calculated using Equation 2.
The amount of variance between the differential voltage present at the device input relative to the voltage developed at the shunt resistor is based both on the external series resistance value as well as the internal input resistors, RINT; see Figure 33. The reduction of the shunt voltage reaching the device input pins appears as a gain error when comparing the output voltage relative to the voltage across the shunt resistor. A factor can be calculated to determine the amount of gain error that is introduced by the addition of external series resistance. Equation 2 calculates the expected deviation from the shunt voltage compared to the expected voltage at the device input pins.
With the adjustment factor equation including the device internal input resistance, this factor varies with each gain version; see Table 1. Each individual device gain error factor is listed in Table 2.
The gain error that can be expected from the addition of the external series resistors can then be calculated based on Equation 3.
|INA250A1||200 mV/A||50 kΩ|
|INA250A2||500 mV/A||20 kΩ|
|INA250A3||800 mV/A||12.5 kΩ|
|INA250A4||2 V/A||5 kΩ|
|DEVICE||SIMPLIFIED GAIN ERROR FACTOR|
For example, using an INA250A2 and the corresponding gain error equation from Table 2, a series resistance of 10 Ω results in a gain error factor of 0.991. The corresponding gain error is then calculated using Equation 3, resulting in a gain error of approximately 0.84% because of the external 10-Ω series resistors.
Although the device does not have a shutdown pin, the low power consumption allows for the device to be powered from the output of a logic gate or transistor switch that can turn on and turn off the voltage connected to the device power-supply pin. However, in current-shunt monitoring applications, there is also a concern for how much current is drained from the shunt circuit in shutdown conditions. Evaluating this current drain involves considering the device simplified schematic in shutdown mode, as shown in Figure 35.
Note that there is typically an approximate 1-MΩ impedance (from the combination of the feedback and input resistors) from each device input to the REF pin. The amount of current flowing through these pins depends on the respective configuration. For example, if the REF pin is grounded, calculating the effect of the 1-MΩ impedance from the shunt to ground is straightforward. However, if the reference or op amp is powered when the device is shut down, the calculation is direct. Instead of assuming 1 MΩ to ground, assume 1 MΩ to the reference voltage. If the reference or op amp is also shut down, some knowledge of the reference or op amp output impedance under shutdown conditions is required. For instance, if the reference source functions similar to an open circuit when un-powered, little or no current flows through the 1-MΩ path.
With a small amount of additional circuitry, the device can be used in circuits subject to transients higher than
36 V (such as in automotive applications). Use only zener diodes or zener-type transient absorbers (sometimes referred to as transzorbs); any other type of transient absorber has an unacceptable time delay. Start by adding a pair of resistors, as shown in Figure 36, as a working impedance for the zener. Keeping these resistors as small as possible is preferable, most often approximately 10 Ω. This value limits the affect on accuracy with the addition of these external components, as described in the Input Filtering section. Device interconnections between the shunt resistor and amplifier have a current handling limit of 1 A. Using a 10-Ω resistor limits the allowable transient range to 10 V above the zener clamp in order to not damage the device. Larger resistor values can be used in this protection circuit to accommodate a larger transient voltage range, resulting in a larger affect on gain error. Because this circuit limits only short-term transients, many applications are satisfied with a 10-Ω resistor along with conventional zener diodes of the lowest power rating available.