TPS6108x is a highly integrated boost regulator for up to 27-V output. In addition to the on-chip 0.5-A/1.2-A PWM switch and power diode, this IC also builds in an input side isolation switch as shown in the block diagram. One common issue with conventional boost regulator is the conduction path from input to output even when PWM switch is turned off. It creates three problems, inrush current during start up, output leakage voltage under shutdown, and unlimited short circuit current. To address these issues, TPS6108x turns off the isolation switch under shutdown mode and short circuit condition to eliminate any possible current path.
TPS6108x adopts current mode control with constant PWM (pulse width modulation) frequency. The switching frequency can be configured to either 600 kHz or 1.2 MHz through the FSW pin. 600 kHz improves light load efficiency, while 1.2 MHz allows using smaller external component. The PWM operation turns on the PWM switch at the beginning of each switching cycle. The input voltage is applied across the inductor and stores the energy as inductor current ramps up. The load current is provided by the output capacitor. When the inductor current across the threshold set by error amplifier output, the PWM switch is turned off, and the power diode is forward biased. The inductor transfers its stored energy to replenish the output capacitor. This operation repeats in the next switching cycle.
The error amplifier compares the FB pin voltage with an internal reference, and its output determines the duty cycle of the PWM switching. This close loop system requires loop compensation for stable operation. TPS6108x has internal compensation circuitry which accommodates a wide range of input and output voltages. The TPS6108x integrates slope compensation to the current ramp to avoid the sub-harmonic oscillation that is intrinsic to current mode control schemes.
TPS6108x turns on the isolation FET when the EN pin is pulled high, provided that the input voltage is higher than the undervoltage lockout threshold. The Vgs of the isolation FET is clamped to maintain high on-resistance and limits the current to 30mA charging the output capacitor. This feature limits the in-rush current and maximum start up current to 30mA. Once the output capacitor is charged to VIN, the IC removes the Vgs clamp to fully turn on the isolation FET and at the same time actives soft start by charging the capacitor on the SS pin. If OUT stays lower than VIN-Vsc following a 1.7ms delay after enable is taken high, the IC recognizes a short circuit condition. In this case, the isolation FET turns off, and IC remains off until the EN pin toggles or VIN cycles through power on reset (POR).
During the soft start phase, the SS pin capacitor is charged by internal bias current of the SS pin. The SS pin capacitor programs the ramp up slope. The SS pin voltage clamps the reference voltage of the FB pin, therefore the output capacitor rise time follows the SS pin voltage. Without the soft start, the inductor current could reach the overcurrent limit threshold, and there is potential for output overshoot. see the Application and Implementation section on selecting soft start capacitor values. Pulling the SS pin to ground disables the PWM switching. However, unlike being disabled by pulling EN low, the IC continues to draw quiescent current and the isolation FET remains on.
TPS6108x has a pulse by pulse overcurrent limit feature which turns off the power switch once the inductor current reaches the overcurrent limit. The PWM circuitry resets itself at the beginning of the next switch cycle. The overcurrent threshold determines the available output current. However, the maximum output is also a function of the input voltage, output voltage, switching frequency and inductor value. Larger inductor values and 1.2MHz switching frequency increase the current output capability because of the reduced current ripple. See the APPLICATION INFORMATION section for the maximum output current calculation.
In typical boost converter topologies, if the output is grounded, turning off the power switch does not limit the current because a current path exists from the input to output through the inductor and power diode. To eliminate this path, TPS6108x turns off the isolation FET between the input and the inductor. This circuit is triggered when the inductor current remains above short circuit current limit for more than 13μs, or the OUT pin voltage falls below VIN-1.4V for more than 1.7ms. An internal catch-diode between the L pin and ground turns on to provide a current discharge path for the inductor. If the short is caused by the output being low, then the IC shuts down and waits for EN to be toggled or a POR. If the short protection is triggered by short circuit current limit, the IC attempts to start up one time. After 57ms, the IC restarts in a fashion described in the above section. If the short is cleared, the boost regulator properly starts up and reaches output regulation. However, after reaching regulation, if another event of short circuit current limit occurs, the IC goes into shutdown mode again, and the fault can only be cleared by toggling the EN pin or POR. Under a permanent short circuit, the IC shuts down after a start up failure and waits for POR or the EN pin toggling.
The same circuit also protects the ICs and external components when the SW pin is shorted to ground. These features provide much more comprehensive and reliable protection than the conventional boost regulator. Table 2 lists the IC protection against the short of each IC pin.
|SHORTED TO GND||FAULT DETECTION||IC OPERATION||HOW TO CLEAR THE FAULT|
|L, SW||INDUCTOR > ISC for 13 μs||Turn off isolation FET||IC restarts after 57ms; If it happens again, the fault can only be cleared by toggling EN or POR.|
|OUT (during start up)||OUT <Vin– 1.4V for 2 ms||IC shuts down||Cleared by toggling EN or POR|
|OUT (after start up)||OUT <Vin– 1.4V without delay||IC shuts down||Cleared by toggling EN or POR|
|FSW||N/A||600 kHz switching frequency||N/A|
|SS||N/A||Disable PWM switching and no output; but still dissipate quiescent current.||N/A|
|FB||N/A||Over voltage protection of the OUT pin||OUT voltage fails by OVP hysteresis|
|GND, PGND, VIN||N/A||N/A||N/A|
When TPS6108x is configured as regulated current output as shown in the Typical Application section, the output voltage can run away if the current load is disconnected. The over voltage condition can also occur if the FB pin is shorted to the ground. To prevent the SW node and the output capacitor from exceeding the maximum voltage rating, an over voltage protection circuit turns off the boost regulator as soon as the output voltage exceeds the OVP threshold. When the output voltage falls 0.7 V below the OVP threshold, the regulator resumes the PWM switching unless the output voltage exceeds the OVP threshold.
An undervoltage lockout prevents mis-operation of the device for input voltages below 1.65 V (typical). When the input voltage is below the undervoltage threshold, the device remains off and both PWM and isolation switch are turned off, providing isolation between input and output. The undervoltage lockout threshold is set below minimum operating voltage of 2.5 V to avoid any transient VIN dip to trigger UVLO and causes converter reset. For the VIN voltage between UVLO threshold and 2.5 V, the IC still maintains its operation. However, the spec is not assured.
An internal thermal shutdown turns off the isolation and PWM switches when the typical junction temperature of 160°C is exceeded. The IC restarts if the junction temperature drops by 15°C.
Connecting the EN pin low turns off the power switch immediately, but keeps the isolation FET on. If the EN pin is logic low for more than 74 ms, the IC turns off the isolation FET and enters shutdown mode drawing less than 1 μA current. The enable input pin has an internal 800 kΩ pulldown resistor to disable the device when the pin is floating.
The FSW pin can be connected to either a logic high or logic low to program the switching frequency to1.2 MHz or 600 kHz respectively. The 600 kHz switching frequency provides better efficiency because of lower switching losses. This advantage becomes more evident at light load when switching losses dominate overall losses. The higher switching frequency shrinks external component size and thus the size of power solution. High switching frequency also improves load transient response because the smaller value inductor takes less time to ramp up and down current. The other benefits of high switching frequency are lower output ripples and a higher maximum output current. Overall, it is recommended to use 1.2 MHz switching frequency unless light load efficiency is a major concern.
The FSW pin has internal 800 kΩ pullup resistor to the VIN pin. Floating this pin programs the switching frequency to 1.2MHz.
The overcurrent limit in a boost converter limits the maximum input current and thus maximum input power from a given input voltage. Maximum output power is less than maximum input power due to power conversion losses. Therefore, the overcurrent limit, the input voltage, the output voltage and the conversion efficiency all affect maximum current output. Because the overcurrent limit clamps the peak inductor current, the current ripple must be subtracted to derive maximum DC current. The current ripple is a function of the switching frequency, the inductor value and the duty cycle.
Ip = inductor peak to peak ripple
L = inductor value
Vf = power diode forward voltage
Fs = Switching frequency
The following equations take into account of all the above factors for maximum output current calculation.
Ilim = overcurrent limit
η = conversion efficiency
To minimize the variation in the overcurrent limit threshold, the TPS6108x uses the VIN and OUT pin voltage to compensate for the variation caused by the slope compensation. However, the threshold still has some dependency on the VIN and OUT voltage. Use Figure 5 to Figure 8 to identify the typical overcurrent limit in your application, and use 25% tolerance to account for temperature dependency and process variations.
Because of the minimum duty cycle of each power switching cycle of TPS6108x, the device can lose regulation at the very light load. Use the following equations to calculate PWM duty cycle under discontinues conduction mode (DCM).
Ipeak = inductor peak to peak ripple in DCM
Iload = load current
D = PWM switching duty cycle
If the calculated duty cycle is less than 5%, minimum load should be considered to the boost output to ensure regulation. Figure 20 provides quick reference to identify the minimum load requirements for two input voltages.