SLUSA55B October   2010  – April 2015

PRODUCTION DATA.  

  1. Features
  2. Applications
  3. Description
  4. Revision History
  5. Device Comparison Table
  6. Pin Configuration and Functions
  7. Specifications
    1. 7.1 Absolute Maximum Ratings
    2. 7.2 ESD Ratings
    3. 7.3 Recommended Operating Conditions
    4. 7.4 Thermal Information
    5. 7.5 Electrical Characteristics
    6. 7.6 Typical Characteristics
  8. Detailed Description
    1. 8.1 Overview
    2. 8.2 Functional Block Diagram
    3. 8.3 Feature Description
      1. 8.3.1  Battery Voltage Regulation
      2. 8.3.2  Battery Current Regulation
      3. 8.3.3  Input Adapter Current Regulation
      4. 8.3.4  Precharge
      5. 8.3.5  Charge Termination, Recharge, and Safety Timer
      6. 8.3.6  Power Up
      7. 8.3.7  Enable and Disable Charging
      8. 8.3.8  System Power Selector
      9. 8.3.9  Automatic Internal Soft-Start Charger Current
      10. 8.3.10 Converter Operation
      11. 8.3.11 Synchronous and Nonsynchronous Operation
      12. 8.3.12 Cycle-by-Cycle Charge Undercurrent Protection
      13. 8.3.13 Input Overvoltage Protection (ACOV)
      14. 8.3.14 Input Undervoltage Lockout (UVLO)
      15. 8.3.15 Battery Overvoltage Protection
      16. 8.3.16 Cycle-by-Cycle Charge Overcurrent Protection
      17. 8.3.17 Thermal Shutdown Protection
      18. 8.3.18 Temperature Qualification
      19. 8.3.19 Timer Fault Recovery
      20. 8.3.20 PG Output
      21. 8.3.21 CE (Charge Enable)
      22. 8.3.22 Charge Status Outputs
      23. 8.3.23 Battery Detection
    4. 8.4 Device Functional Modes
  9. Application and Implementation
    1. 9.1 Application Information
    2. 9.2 Typical Application
      1. 9.2.1 Design Requirements
      2. 9.2.2 Detailed Design Procedure
        1. 9.2.2.1 Inductor Selection
        2. 9.2.2.2 Input Capacitor
        3. 9.2.2.3 Output Capacitor
        4. 9.2.2.4 Power MOSFET Selection
        5. 9.2.2.5 Input Filter Design
        6. 9.2.2.6 Inductor, Capacitor, and Sense Resistor Selection Guidelines
        7. 9.2.2.7 Component List for Typical System Circuit of
      3. 9.2.3 Application Curves
  10. 10Power Supply Recommendations
  11. 11Layout
    1. 11.1 Layout Guidelines
    2. 11.2 Layout Example
  12. 12Device and Documentation Support
    1. 12.1 Device Support
      1. 12.1.1 Third-Party Products Disclaimer
    2. 12.2 Documentation Support
      1. 12.2.1 Related Documentation
    3. 12.3 Trademarks
    4. 12.4 Electrostatic Discharge Caution
    5. 12.5 Glossary
  13. 13Mechanical, Packaging, and Orderable Information

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サーマルパッド・メカニカル・データ
発注情報

9 Application and Implementation

NOTE

Information in the following applications sections is not part of the TI component specification, and TI does not warrant its accuracy or completeness. TI’s customers are responsible for determining suitability of components for their purposes. Customers should validate and test their design implementation to confirm system functionality.

9.1 Application Information

The bq24618 battery charger is ideal for high current charging (up to 10 A) and can charge battery packs consisting of single cells or multiple cells in series. The bq24610EVM evaluation module is a complete charge module for evaluating the bq2461x. The application curves were taken using the bq24610EVM. Refer to the EVM user's guide (SLUU396) for EVM information.

9.2 Typical Application

bq24618 typ_sch_lusa55.gif
VIN = 19 V, 3-cell, Iadapter_limit = 4 A, Icharge = 3 A, Ipre-charge = Iterm = 0.3 A, 5-hour safety timer
Figure 19. Typical System Schematic

9.2.1 Design Requirements

For this design example, use the parameters listed in Table 3 as the input parameters.

Table 3. Design Parameters

DESIGN PARAMETER VALUE
AC adapter voltage (VIN) 19 V
AC adapter current limit 4 A
Battery charge voltage (number of cells in series) 12.6 V (3 cells)
Battery charge current (during constant current phase) 3 A
Precharge and termination current 0.3 A
Safety timer 5 hours

9.2.2 Detailed Design Procedure

9.2.2.1 Inductor Selection

The bq2461x has 600-kHz switching frequency to allow the use of small inductor and capacitor values. Inductor saturation current should be higher than the charging current (ICHG) plus half the ripple current (IRIPPLE):

Equation 12. bq24618 EQ6_Isat_lus875.gif

The inductor ripple current depends on input voltage (VIN), duty cycle (D = VOUT/VIN), switching frequency (fS) and inductance (L):

Equation 13. bq24618 EQ7_Iripp_lus875.gif

The maximum inductor ripple current happens with D = 0.5 or close to 0.5. For example, the battery charging voltage range is from 9 V to 12.6 V for a three-cell battery pack. For 20-V adapter voltage, 10-V battery voltage gives the maximum inductor ripple current. Another example is a four-cell battery, where the battery voltage range is from 12 V to 16.8 V, and 12-V battery voltage gives the maximum inductor ripple current.

Usually inductor ripple is designed in the range of 20% to 40% of maximum charging current as a trade-off between inductor size and efficiency for a practical design.

The bq24618 has cycle-by-cycle charge undercurrent protection (UCP) by monitoring the charge current-sensing resistor to prevent negative inductor current. The typical UCP threshold is 5 mV falling edge, corresponding to 0.5 A falling edge for a 10-mΩ charge current-sensing resistor.

9.2.2.2 Input Capacitor

The input capacitor should have enough ripple current rating to absorb the input switching ripple current. The worst-case RMS ripple current is half of the charging current when the duty cycle is 0.5. If the converter does not operate at 50% duty cycle, then the worst-case capacitor RMS current ICIN occurs where the duty cycle is closest to 50% and can be estimated by the following equation:

Equation 14. bq24618 EQ8_Icin_lus875.gif

A low-ESR ceramic capacitor such as X7R or X5R is preferred for the input decoupling capacitor and should be placed as close as possible to the drain of the high-side MOSFET and source of the low-side MOSFET. The voltage rating of the capacitor must be higher than the normal input voltage level. A 25-V rating or higher capacitor is preferred for 20-V input voltage. A 10-µF to 20-µF capacitor is suggested for typical 3-A to 4-A charging current.

9.2.2.3 Output Capacitor

The output capacitor also should have enough ripple current rating to absorb output switching ripple current. The output capacitor RMS current ICOUT is given:

Equation 15. bq24618 EQ9_Icout_lus875.gif

The output capacitor voltage ripple can be calculated as follows:

Equation 16. bq24618 eqad1a_vo_lus892.gif

At certain input/output voltage and switching frequency, the voltage ripple can be reduced by increasing the output filter LC.

The bq24618 has an internal loop compensator. To get good loop stability, the resonant frequency of the output inductor and output capacitor should be designed from 12 kHz to 17 kHz. The preferred ceramic capacitor is 25-V or higher rating, X7R or X5R for 4-cell applications.

9.2.2.4 Power MOSFET Selection

Two external N-channel MOSFETs are used for a synchronous switching battery charger. The gate drivers are internally integrated into the IC with 6 V of gate drive voltage. MOSFETs of 30-V or higher voltage rating are preferred for 20-V input voltage and 40-V or higher rating MOSFETs are preferred for 20-V to 28-V input voltage.

Figure-of-merit (FOM) is usually used for selecting proper the MOSFET, based on a trade-off between the conduction loss and switching loss. For a top-side MOSFET, FOM is defined as the product of a MOSFET ON-resistance, rDS(on), and the gate-to-drain charge, QGD. For a bottom-side MOSFET, FOM is defined as the product of the MOSFET ON-resistance, rDS(on), and the total gate charge, QG.

Equation 17. bq24618 EQ10_FOM_lus875.gif

The lower the FOM value, the lower the total power loss. Usually, lower rDS(on) has higher cost with the same package size.

The top-side MOSFET loss includes conduction loss and switching loss. It is a function of duty cycle (D = VOUT/VIN), charging current (ICHG), MOSFET ON-resistance rDS(on)), input voltage (VIN), switching frequency (fS), turnon time (ton) and turnoff time (toff):

Equation 18. bq24618 EQ11_Ptop_lus875.gif

The first item represents the conduction loss. Usually MOSFET rDS(on) increases by 50% with a 100ºC junction temperature rise. The second term represents the switching loss. The MOSFET turnon and turnoff times are given by:

Equation 19. bq24618 EQ12_ton_lus875.gif

where

  • Qsw is the switching charge.
  • Ion is the turnon gate-drive current.
  • Ioff is the turnoff gate-drive current.

If the switching charge is not given in the MOSFET data sheet, it can be estimated by gate-to-drain charge (QGD) and gate-to-source charge (QGS):

Equation 20. bq24618 EQ13_QSW_lus875.gif

Total gate drive current can be estimated by the REGN voltage (VREGN), MOSFET plateau voltage (Vplt), total turnon gate resistance (Ron), and turnoff gate resistance (Roff) of the gate driver:

Equation 21. bq24618 EQ14_Ion_lus875.gif

The conduction loss of the bottom-side MOSFET is calculated with the following equation when it operates in synchronous continuous conduction mode:

Equation 22. bq24618 EQ15_Pbott_lus875.gif

If the SRP-SRN voltage decreases below 5 mV (the charger is also forced into nonsynchronous mode when the average SRP-SRN voltage is lower than 1.25 mV), the low-side FET is turned off for the remainder of the switching cycle to prevent negative inductor current.

As a result, all the freewheeling current goes through the body diode of the bottom-side MOSFET. The maximum charging current in nonsynchronous mode can be up to 0.9 A (0.5 A typical) for a 10-mΩ charging current sensing resistor, considering IC tolerance. Choose the bottom-side MOSFET with either an internal Schottky or body diode capable of carrying the maximum nonsynchronous mode charging current.

MOSFET gate driver power loss contributes to the dominant losses on the controller IC when the buck converter is switching. Choosing a MOSFET with a small Qg_total reduces the IC power loss to avoid thermal shutdown.

Equation 23. bq24618 eqad2_IC_lus892.gif

where

  • Qg_total is the total gate charge for both upper and lower MOSFET at 6 V VREGN.

9.2.2.5 Input Filter Design

During adapter hot plug-in, the parasitic inductance and input capacitor from the adapter cable form a second-order system. The voltage spike at the VCC pin may be beyond the IC maximum voltage rating and damage the IC. The input filter must be carefully designed and tested to prevent an overvoltage event on the VCC pin. The ACP/ACN pin must be placed after the input ACFET in order to avoid overvoltage stress on these pins during hot plug-in.

There are several methods to damping or limiting the overvoltage spike during adapter hot plug-in. An electrolytic capacitor with high ESR as an input capacitor can damp the overvoltage spike well below the IC maximum pin voltage rating. A high current capability TVS Zener diode can also limit the overvoltage level to an IC-safe level. However, these two solutions may not have low cost or small size.

A cost-effective and small-size solution is shown in Figure 20. R1 and C1 comprise a damping RC network to damp the hot plug-in oscillation. As a result, the overvoltage spike is limited to a safe level. D1 is used for reverse voltage protection for the VCC pin (it can be the body diode of the input ACFET). C2 is a VCC pin decoupling capacitor, and it should be placed as close as possible to the VCC pin. R2 and C2 form a damping RC network to further protect the IC from high dv/dt and high-voltage spikes. The value of C2 should be less than the value of C1 so R1 can be dominant over the ESR orf C1 to get enough damping effect for hot plug-in. The R1 and R2 packages must be sized to handle the inrush current power loss according to the resistor manufacturer’s data sheet. The filter component values always must be verified with the real application, and minor adjustments may be needed to fit in the real application circuit.

bq24618 IP_flt_lus8892.gifFigure 20. Input Filter

9.2.2.6 Inductor, Capacitor, and Sense Resistor Selection Guidelines

The bq24618 provides internal loop compensation. With this scheme, best stability occurs when the LC resonant frequency, fo, is approximately 12 kHz to 17 kHz for the bq24618.

The following table provides a summary of typical LC components for various charge currents:

Table 4. Typical Inductor, Capacitor, and Sense Resistor Values as a Function of Charge Current for bq24618 (600-kHz Switching Frequency)

CHARGE CURRENT 2 A 4 A 6 A 8 A 10 A
Output inductor LO 6.8 μH 6.8 μH 4.7 μH 3.3 μH 3.3 μH
Output capacitor CO 20 μF 20 μF 30 μF 40 μF 40 μF
Sense resistor 10 mΩ 10 mΩ 10 mΩ 10 mΩ 10 mΩ

9.2.2.7 Component List for Typical System Circuit of Figure 19

PART DESIGNATOR QTY DESCRIPTION
Q1, Q2, Q3 2 P-channel MOSFET, –30 V, –35 A, PowerPAK 1212-8, Vishay-Siliconix, Si7617DN
Q4, Q5 2 N-channel MOSFET, 30 V, 12 A, PowerPAK 1212-8, Vishay-Siliconix, Sis412DN
D1 1 Diode, dual Schottky, 30 V, 200 mA, SOT23, Fairchild, BAT54C
D2, D3, D4 3 LED diode, green, 2.1 V, 20 mA, LTST-C190GKT
RAC, RSR 2 Sense resistor, 10 mΩ, 2010, Vishay-Dale, WSL2010R0100F
L1 1 Inductor, 6.8 µH, 5.5 A, Vishay-Dale IHLP2525CZ
C8, C9, C12, C13 4 Capacitor, ceramic, 10 µF, 35 V, 20%, X7R
C4, C5 2 Capacitor, ceramic, 1 µF, 16 V, 10%, X7R
C1, C3, C6, C11 4 Capacitor, ceramic, 0.1 µF, 16 V, 10%, X7R
C2, C10 2 Capacitor, ceramic, 0.1 µF, 50 V, 10%, X7R
C7 1 Capacitor, ceramic, 1 µF, 50 V, 10%, X7R
C14, C15 (Optional) 2 Capacitor, ceramic, 0.1 µF, 50 V, 10%, X7R
C16 1 Capacitor, ceramic, 2.2 µF, 35 V, 10%, X7R
Cff 1 Capacitor, ceramic, 22 pF, 25 V, 10%, X7R
CTTC 1 Capacitor, ceramic, 0.056 µF, 16 V, 5%, X7R
R1, R3, R5, R7 4 Resistor, chip, 100 kΩ, 1/16 W, 0.5%
R2 1 Resistor, chip, 500 kΩ, 1/16 W, 0.5%
R4 1 Resistor, chip, 32.4 kΩ, 1/16 W, 0.5%
R6 1 Resistor, chip, 10 kΩ, 1/16 W, 0.5%
R8 1 Resistor, chip, 22.1 kΩ, 1/16 W, 0.5%
R9 1 Resistor, chip, 9.31 kΩ, 1/16 W, 1%
R10 1 Resistor, chip, 430 kΩ, 1/16 W, 1%
R11, R12, R13, R18, R19 5 Resistor, chip, 10 kΩ, 1/16 W, 5%
R14, R15 (optional) 2 Resistor, chip, 100 kΩ, 1/16 W, 5%
R16 1 Resistor, chip, 100 Ω, 1/16 W, 5%
R17 1 Resistor, chip, 10 Ω, 1/4 W, 5%
R20 1 Resistor, chip, 2 Ω, 1 W, 5%

9.2.3 Application Curves

bq24618 Fig_20.png
VIN: 19 V VBAT: 12 V ICHG = 4 A
Figure 21. Continuous Conduction Mode Switching Waveform
bq24618 Fig_21.png
VIN: 19 V VBAT: 12 V
Figure 22. Battery Charging Soft Start