Information in the following applications sections is not part of the TI component specification, and TI does not warrant its accuracy or completeness. TI’s customers are responsible for determining suitability of components for their purposes. Customers should validate and test their design implementation to confirm system functionality.
The TPS6104x is designed for output voltages up to 28 V with an input voltage range of 1.8 V to 6 V and a switch peak current limit of 400 mA (250 mA for the TPS61041). The device operates in a pulse-frequency-modulation (PFM) scheme with constant peak current control. This control scheme maintains high efficiency over the entire load current range, and with a switching frequency up to 1 MHz, the device enables the use of very small external components. The following section provides a step-by-step design approach for configuring the TPS61040 as a voltage regulating boost converter for LCD bias power supply, as shown in Figure 12.
The following section provides a step-by-step design approach for configuring the TPS611040 as a voltage regulating boost converter for LCD bias supply, as shown in Figure 12.
|DESIGN PARAMETER||EXAMPLE VALUE|
|Input Voltage||1.8 V to 6 V|
|Output Voltage||18 V|
|Output Current||10 mA|
Because the PFM peak current control scheme is inherently stable, the inductor value does not affect the stability of the regulator. The selection of the inductor together with the nominal load current, input and output voltage of the application determines the switching frequency of the converter. Depending on the application, inductor values from 2.2 μH to 47 μH are recommended. The maximum inductor value is determined by the maximum on time of the switch, typically 6 μs. The peak current limit of 400 mA/250 mA (typically) should be reached within this 6-μs period for proper operation.
The inductor value determines the maximum switching frequency of the converter. Therefore, select the inductor value that ensures the maximum switching frequency at the converter maximum load current is not exceeded. The maximum switching frequency is calculated by the following formula:
If the selected inductor value does not exceed the maximum switching frequency of the converter, the next step is to calculate the switching frequency at the nominal load current using the following formula:
A smaller inductor value gives a higher converter switching frequency, but lowers the efficiency.
The inductor value has less effect on the maximum available load current and is only of secondary order. The best way to calculate the maximum available load current under certain operating conditions is to estimate the expected converter efficiency at the maximum load current. This number can be taken out of the efficiency graphs shown in Figure 1 through Figure 4. The maximum load current can then be estimated as follows:
The maximum load current of the converter is the current at the operation point where the converter starts to enter the continuous conduction mode. Usually the converter should always operate in discontinuous conduction mode.
Last, the selected inductor should have a saturation current that meets the maximum peak current of the converter (as calculated in Peak Current Control). Use the maximum value for ILIM for this calculation.
Another important inductor parameter is the dc resistance. The lower the dc resistance, the higher the efficiency of the converter. See Table 3 and the typical applications for the inductor selection.
|DEVICE||INDUCTOR VALUE||COMPONENT SUPPLIER||COMMENTS|
|TPS61040||10 μH||Sumida CR32-100||High efficiency|
|10 μH||Sumida CDRH3D16-100||High efficiency|
|10 μH||Murata LQH4C100K04||High efficiency|
|4.7 μH||Sumida CDRH3D16-4R7||Small solution size|
|4.7 μH||Murata LQH3C4R7M24||Small solution size|
|TPS61041||10 μH||Murata LQH3C100K24||High efficiency
Small solution size
The output voltage is calculated as:
For battery-powered applications, a high-impedance voltage divider should be used with a typical value for R2 of ≤200 kΩ and a maximum value for R1 of 2.2 MΩ. Smaller values might be used to reduce the noise sensitivity of the feedback pin.
A feedforward capacitor across the upper feedback resistor R1 is required to provide sufficient overdrive for the error comparator. Without a feedforward capacitor, or one whose value is too small, the TPS6104x shows double pulses or a pulse burst instead of single pulses at the switch node (SW), causing higher output voltage ripple. If this higher output voltage ripple is acceptable, the feedforward capacitor can be left out.
The lower the switching frequency of the converter, the larger the feedforward capacitor value required. A good starting point is to use a 10-pF feedforward capacitor. As a first estimation, the required value for the feedforward capacitor at the operation point can also be calculated using the following formula:
The larger the feedforward capacitor the worse the line regulation of the device. Therefore, when concern for line regulation is paramount, the selected feedforward capacitor should be as small as possible. See the following section for more information about line and load regulation.
The line regulation of the TPS6104x depends on the voltage ripple on the feedback pin. Usually a 50 mV peak-to-peak voltage ripple on the feedback pin FB gives good results.
Some applications require a very tight line regulation and can only allow a small change in output voltage over a certain input voltage range. If no feedforward capacitor CFF is used across the upper resistor of the voltage feedback divider, the device has the best line regulation. Without the feedforward capacitor the output voltage ripple is higher because the TPS6104x shows output voltage bursts instead of single pulses on the switch pin (SW), increasing the output voltage ripple. Increasing the output capacitor value reduces the output voltage ripple.
If a larger output capacitor value is not an option, a feedforward capacitor CFF can be used as described in the previous section. The use of a feedforward capacitor increases the amount of voltage ripple present on the feedback pin (FB). The greater the voltage ripple on the feedback pin (≥50 mV), the worse the line regulation. There are two ways to improve the line regulation further:
For best output voltage filtering, a low ESR output capacitor is recommended. Ceramic capacitors have a low ESR value but tantalum capacitors can be used as well, depending on the application.
Assuming the converter does not show double pulses or pulse bursts on the switch node (SW), the output voltage ripple can be calculated as:
|DEVICE||CAPACITOR||VOLTAGE RATING||COMPONENT SUPPLIER(1)||COMMENTS|
|TPS6104x||4.7 μF/X5R/0805||6.3 V||Tayo Yuden JMK212BY475MG||CIN/COUT|
|10 μF/X5R/0805||6.3 V||Tayo Yuden JMK212BJ106MG||CIN/COUT|
|1 μF/X7R/1206||25 V||Tayo Yuden TMK316BJ105KL||COUT|
|1 μF/X5R/1206||35 V||Tayo Yuden GMK316BJ105KL||COUT|
|4.7 μF/X5R/1210||25 V||Tayo Yuden TMK325BJ475MG||COUT|
For good input voltage filtering, low ESR ceramic capacitors are recommended. A 4.7-μF ceramic input capacitor is sufficient for most of the applications. For better input voltage filtering this value can be increased. See Table 4 and typical applications for input capacitor recommendations.
To achieve high efficiency a Schottky diode should be used. The current rating of the diode should meet the peak current rating of the converter as it is calculated in Peak Current Control. Use the maximum value for ILIM for this calculation. See Table 5 and the typical applications for the selection of the Schottky diode.
|DEVICE||REVERSE VOLTAGE||COMPONENT SUPPLIER(1)||COMMENTS|
|TPS6104x||30 V||ON Semiconductor MBR0530|
|20 V||ON Semiconductor MBR0520|
|20 V||ON Semiconductor MBRM120L||High efficiency|
|30 V||Toshiba CRS02|