8.1 Application Information

These are op amps with JFET input devices. These JFETs have large reverse breakdown voltages from gate to source and drain eliminating the need for clamps across the inputs. Therefore large differential input voltages can easily be accommodated without a large increase in input current. The maximum differential input voltage is independent of the supply voltages. However, neither of the input voltages should be allowed to exceed the negative supply as this will cause large currents to flow which can result in a destroyed unit.

Exceeding the negative common-mode limit on either input will force the output to a high state, potentially causing a reversal of phase to the output. Exceeding the negative common-mode limit on both inputs will force the amplifier output to a high state. In neither case does a latch occur since raising the input back within the common-mode range again puts the input stage and thus the amplifier in a normal operating mode.

Exceeding the positive common-mode limit on a single input will not change the phase of the output however, if both inputs exceed the limit, the output of the amplifier will be forced to a high state.

These amplifiers will operate with the common-mode input voltage equal to the positive supply. In fact, the common-mode voltage can exceed the positive supply by approximately 100 mV independent of supply voltage and over the full operating temperature range. The positive supply can therefore be used as a reference on an input as, for example, in a supply current monitor and/or limiter.

Precautions should be taken to ensure that the power supply for the integrated circuit never becomes reversed in polarity or that the unit is not inadvertently installed backwards in a socket as an unlimited current surge through the resulting forward diode within the IC could cause fusing of the internal conductors and result in a destroyed unit.

All of the bias currents in these amplifiers are set by FET current sources. The drain currents for the amplifiers are therefore essentially independent of supply voltage.

As with most amplifiers, care should be taken with lead dress, component placement and supply decoupling in order to ensure stability. For example, resistors from the output to an input should be placed with the body close to the input to minimize pick-up and maximize the frequency of the feedback pole by minimizing the capacitance from the input to ground.

A feedback pole is created when the feedback around any amplifier is resistive. The parallel resistance and capacitance from the input of the device (usually the inverting input) to AC ground set the frequency of the pole. In many instances the frequency of this pole is much greater than the expected 3-dB frequency of the closed loop gain and consequently there is negligible effect on stability margin. However, if the feedback pole is less than approximately six times the expected 3-dB frequency a lead capacitor should be placed from the output to the input of the op amp. The value of the added capacitor should be such that the RC time constant of this capacitor and the resistance it parallels is greater than or equal to the original feedback pole time constant.

8.2 Typical Application

Figure 35. Settling Time Test Circuit

8.2.3 Application Curves

Large Signal Inverter Output, V_OUT (from Settling Time Circuit)

Figure 36. LF355

Figure 38. LF357

Figure 37. LF356

8.3 System Examples

Figure 39. Low Drift Adjustable Voltage Reference

• ΔV_OUT / ΔT = ±0.002%/°C
• All resistors and potentiometers should be wire-wound
• P1: drift adjust
• P2: V_OUT adjust
• Use LF155 for
  • Low I_B
  • Low drift
  • Low supply current

Figure 40. Fast Logarithmic Converter

• Dynamic range: 100 μA ≤ I_i ≤ 1 mA (5 decades), |V_O| = 1 V/decade
• Transient response: 3 μs for ΔI_i = 1 decade
• C1, C2, R2, R3: added dynamic compensation
• V_OS adjust the LF156 to minimize quiescent error
• R_T: Tel Labs type Q81 + 0.3%/°C
Equation 1.

Figure 41. Precision Current Monitor

• V_O = 5 R1/R2 (V/mA of I_S)
• R1, R2, R3: 0.1% resistors
• Use LF155 for
  • Common-mode range to supply range
  • Low I_B
  • Low V_OS
  • Low Supply Current

Figure 42. 8-Bit D/A Converter With Symmetrical Offset Binary Operation

• R1, R2 should be matched within ±0.05%
• Full-scale response time: 3 μs

Figure 43. Wide BW Low Noise, Low Drift Amplifier

Equation 2.

Parasitic input capacitance C1 ≃ (3 pF for LF155, LF156 and LF357 plus any additional layout capacitance) interacts with feedback elements and creates undesirable high frequency pole. To compensate add C2 such that: R2 C2 ≃ R1 C1.

Figure 44. Boosting the LF156 With a Current Amplifier

• I_OUT(MAX) ≃ 150 mA (will drive R_L ≥ 100 Ω)
Equation 3.
• No additional phase shift added by the current amplifier

Figure 45. Decades VCO

R1, R4 matched. Linearity 0.1% over 2 decades.

Equation 4.

Figure 46. Isolating Large Capacitive Loads

• Overshoot 6%
• t_s 10 μs
• When driving large C_L, the V_OUT slew rate determined by C_L and I_OUT(MAX):
Equation 5.

Figure 47. Low Drift Peak Detector

• By adding D1 and R_f, V_D1 = 0 during hold mode. Leakage of D2 provided by feedback path through R_f.
• Leakage of circuit is essentially I_b (LF155, LF156) plus capacitor leakage of Cp.
• Diode D3 clamps V_OUT (A1) to V_IN − V_D3 to improve speed and to limit reverse bias of D2.
• Maximum input frequency should be << ½πR_fC_D2 where C_D2 is the shunt capacitance of D2.

Figure 48. Noninverting Unity Gain Operation for LF157

Equation 6.

Figure 49. Inverting Unity Gain for LF157

Equation 7.

Figure 50. High Impedance, Low Drift Instrumentation Amplifier

• System V_OS adjusted via A2 V_OS adjust
• Trim R3 to boost up CMRR to 120 dB. Instrumentation amplifier resistor array recommended for best accuracy and lowest drift
Equation 8.

Figure 51. Fast Sample and Hold

• Both amplifiers (A1, A2) have feedback loops individually closed with stable responses (overshoot negligible)
• Acquisition time T_A, estimated by:
Equation 9.
• LF156 develops full S_r output capability for V_IN ≥ 1 V
• Addition of SW2 improves accuracy by putting the voltage drop across SW1 inside the feedback loop
• Overall accuracy of system determined by the accuracy of both amplifiers, A1 and A2

Figure 52. High Accuracy Sample and Hold

• By closing the loop through A2, the V_OUT accuracy will be determined uniquely by A1.
  • No V_OS adjust required for A2.
• T_A can be estimated by same considerations as previously but, because of the added
  • propagation delay in the feedback loop (A2) the overshoot is not negligible.
• Overall system slower than fast sample and hold
• R1, C_C: additional compensation
• Use LF156 for
  • Fast settling time
  • Low V_OS

Figure 53. High Q Band Pass Filter

• By adding positive feedback (R2)
• Q increases to 40
• f_BP = 100 kHz
Equation 10.
• Clean layout recommended
• Response to a 1-V_p-p tone burst: 300 μs

Figure 54. High Q Notch Filter

• 2R1 = R = 10 MΩ
  • 2C = C1 = 300 pF
• Capacitors should be matched to obtain high Q
• f_NOTCH = 120 Hz, notch = −55 dB, Q > 100
• Use LF155 for
  • Low I_B
  • Low supply current

Figure 55. V_OS Adjustment

• V_OS is adjusted with a 25-k potentiometer
• The potentiometer wiper is connected to V⁺
• For potentiometers with temperature coefficient of 100 ppm/°C or less the additional drift with adjust
  is ≈ 0.5 μV/°C/mV of adjustment
• Typical overall drift: 5 μV/°C ±(0.5 μV/°C/mV of adj.)

Figure 56. Driving Capacitive Loads

• ^*LF15x R = 5k, LF357 R = 1.25 k
• Due to a unique output stage design, these amplifiers have the ability to drive large capacitive loads and still maintain stability. C_L(MAX) ≃ 0.01 μF.
• Overshoot ≤ 20%, Settling time (t_s) ≃ 5 μs

Figure 57. LF357 - A Large Power BW Amplifier

For distortion ≤ 1% and a 20 Vp-p V_OUT swing, power bandwidth is: 500 kHz.