TIDUF21A December   2022  – January 2023

 

  1.   Description
  2.   Resources
  3.   Features
  4.   Applications
  5.   5
  6. 1System Description
  7. 2System Overview
    1. 2.1 Block Diagram
    2. 2.2 Design Considerations
      1. 2.2.1 Description of Control Logic
      2. 2.2.2 Behavior Throughout Charge Cycle
      3. 2.2.3 Additional Design Recommendations
      4. 2.2.4 Simulation Results
    3. 2.3 Highlighted Products
      1. 2.3.1 TPSI3052-Q1
      2. 2.3.2 TLV7011
      3. 2.3.3 UCC27517A-Q1
  8. 3Hardware, Software, Testing Requirements, and Test Results
    1. 3.1 Hardware Requirements
    2. 3.2 Test Setup
    3. 3.3 Test Results
  9. 4Design and Documentation Support
    1. 4.1 Design Files
      1. 4.1.1 Schematics
      2. 4.1.2 BOM
    2. 4.2 Documentation Support
    3. 4.3 Support Resources
    4. 4.4 Trademarks
  10. 5About the Authors
  11. 6Revision History

2.2.2 Behavior Throughout Charge Cycle

Since no current is flowing through RSHUNT initially, the circuit logic turns ON transistor M1. When transistor M1 is turned ON, the current through the inductor begins rise following constant rate of charge as shown in I of Figure 2-2. Once the current in the inductor has increased such that the voltage across RSHUNT generates a voltage VH and the output of the comparator switches to 0-V. At this point transistor M1 is turned OFF. When this transistor is switched OFF, the current flow through the inductor tries to continue flowing in the same direction. The magnetic field of the inductor will force a negative voltage on the switch node (SW). Once the voltage has reached the forward voltage drop of the free-wheeling diode D1, this diode will turn ON creating a negative voltage across the inductor and the current through the inductor begins to decrease as shown in Figure 2-2. When the current through the inductor creates a voltage across RSHUNT equal to VL, the output of the comparator will go high to 5-V. At this point transistor M1 is turned ON and the switching cycle repeats until the output voltage in the capacitor equals the input voltage. Once the output voltage equals the input voltage no current will flow in the inductor and the transistor M1 will remain on until the precharge circuit is turned off.

Figure 2-2 Charging Plot Behavior

The current through the inductor can be studied by separating it into three different sections I, II, and III. See Figure 2-2. It is necessary to separate the current into these three sections and understand the dependencies because the behavior will directly affect the design requirements. Section I and III have the fastest slew rate. This fast slew rate is due to the high voltage across the inductor. Initially the capacitor does not have any charge and the voltage across the inductor when M1 is ON would be the battery voltage. Slew rate must be carefully taken into account when selecting the comparator and the push-pull driver. The propagation delay of these components will add up to the overall control loop delay.

The following equation describes the behavior for the slew rate. The fastest slew rate occurs at the beginning of the pre-charge cycle, where V is equal to the system voltage (for example, 800-V). Therefore only the inductance value can be manipulated to set the peak slew rate. The higher the inductance the lower the slew rate for the current through the inductor.

Equation 12. dIdt = VL 

For example, if a 800-V system with a 560-uH inductor is designed to have a IPEAK of 8-A, the di/dt is 1.429-A/us, and the loop propagation delay is 1-us. Then the effective IPEAK will be 9.429-A instead of 8-A.

In addition to the slew rate, another important specification is the maximum required switching frequency. Since TPSI3052-Q1 has a limited power transfer, it is required and critical to ensure that TPSI3052-Q1 can achieve the expected switching frequency after providing power to the control loop devices. The following equation describes the relation between the maximum switching frequency seen during the precharge cycle and the design components. There are only two design parameters that can be set to ensure that the switching frequency remains low. Inductance and the current swing dI. At this point it is easy to see that the inductor is a critical components for this design. High inductance value with high current rating would the best options to limit the switching frequency and the slew rate for the system. Once the maximum switching frequency seen during precharge for the design is calculated, use the TPSI3052-Q1 excel calculator to make sure the TPSI3052-Q1 can supply the necessary power at that switching frequency.

Equation 13. FMAX=Vbatt22×L×dI 

Vbatt = battery voltage

L = inductance

dI = inductor peak-to-peak current

Now that all the equations and limitations have been studied and defined with equations, it can be easier to go through the design procedures for this reference design.

Table 2-1 Design Requirements
SpecificationsMaxComments
Battery voltage800-VThe case of 800-V system was used for this design procedure.
Total system capacitance2-mFThis value represents the addition of all the capacitance on the HV DC-link bus.
Required charging time400-ms

Duration of time the capacitor needs to charge from fully discharged, 0-V, up to the battery voltage level, 800-V. A short precharge time is preferred so that the vehicle starts quickly after power on.

Maximum Switching frequency50-kHzThe switching frequency is limited by the maximum power transfer by the TPSI3052-Q1. Dependent on the total gate charge (Qg) of the FET, the inductance and the allowed ripple current.
Figure 2-3 Block Diagram

The first value to calculate for the design is the required average current to precharge the capacitor within the required time.

Equation 14. I=CdVdt = (2 mF)(800V)(400 ms) = 4 A

The average current is important for the charging time requirements, but even more important for this system is the peak-to-peak current (dI). Since the switching frequency must be limited, then dI must be large enough to maintain the switching frequency bounded. For this design the selected IPEAK is 8-A and the IMIN 0.5-A which results in a total dI of 7.5-A. In addition, the inductance value would be another critical parameter to help reduce the switching frequency. The inductor selected for this design has an inductance of 560-uH, a saturation current rating of 8.6-A, a RMS current rating of 4.6-A, and a voltage rating of 1000-V. This inductor satisfy our voltage rating or 800-V, IPEAK 0f 8-A, and IAVG of 4-A. The following equations describe the expected maximum switching frequency for the system and the maximum slew rate.

Equation 15. FMAX=Vbatt22×L×dI = 400 V2(560 uH)(7.5 A) = 47 kHz
Equation 16. dIdt  = VL = 800 V560 uH = 1 A/us

The previous equation describes the fastest current slope. This happens when the pre-charge has just started and the voltage on the capacitor is 0-V. This value is critical because if the control loop has a 1-us propagation delay, then the peak current can be 1-A higher than expected. Once IPEAK and IMIN are determined, then the thresholds for the comparator must be calculated and from there the resistors for the hysteresis. The following equations show how to calculate these values.

Equation 17. VH = IPEAK × RSHUNT = 8 A × 100 mΩ = 800 mV
Equation 18. VL = IMIN × RSHUNT = 0.5 A × 100 mΩ = 50 mV
Equation 19. R2R1 = VLVH - VL = 50 mV800 mV - 50 mV ≈ 0.06666
Equation 20. R3R1 = VL+5V_Viso - VH = 50 mV5 V - 800 mV ≈ 0.0119

If R1 = 200 kΩ, then R2 = 13.3 kΩ, and R3 = 2.38 kΩ.

The final step to complete the design is selecting the FETs and free wheeling diode. For the FETs, it is recommended to select FETs with low gate charge (Qg). Low Qg will allow the TPSI3052-Q1 to achieve higher switching frequency. Visit the following excel calculator to assist calculating the max switching frequency depending on the total Qg.