STDA011 September   2025 UCC25661

 

  1.   1
  2.   Abstract
  3.   Trademarks
  4. 1Introduction
  5. 2Benefits of GaN In LLC Resonant Converter
    1. 2.1 Higher Efficiency
    2. 2.2 Faster Switching Speeds
    3. 2.3 Reduced Parasitic Capacitances
    4. 2.4 Improved Power Density
    5. 2.5 High Thermal Conductivity
    6. 2.6 Lower Junction Temperatures
  6. 3LLC Resonant Converter
    1. 3.1 The Relationship Between Output Voltage (VOUT) and Switching Frequency (fS) in an LLC Resonant Controller
      1. 3.1.1 The LLC Charging Challenge
      2. 3.1.2 A Wide VIN/VOUT Capable LLC
  7. 4Practical Application of LLC Converters for a Battery Charger Leveraging GaN Switches
    1. 4.1 Requirements and Scope
    2. 4.2 Charging Curve for Lithium-Ion Battery
    3. 4.3 How to Support Wide VOUT Range in an LLC Design for Battery Chargers
    4. 4.4 The Prototype Hardware
  8. 5Summary

4.3 How to Support Wide VOUT Range in an LLC Design for Battery Chargers

The first point is to achieve ZVS in the whole Vin / Vout / Iout range, then regulate VOUT in a wide range, suitable for battery chargers.

The challenge is that the magnetizing current changes a lot versus switching frequency, output voltage and input voltage; all these three parameters are connected between each other through the gain – frequency relationship as shown below. Here we highlighted two main areas where the converter works far away from resonance. As explained at the beginning, at fS > fR the gain of the LLC tends to zero only if enough load current is present, typically in overload condition or close to short. For all other conditions (light or mid load) the gain is slightly less than 1. For battery charger it’s more challenging because the battery voltage level is requiring gain variation to achieve values well below 1. In this regard the controller, with a combination of low and high-frequency burst mode, is adapting the equivalent gain to be way lower than 1, without any particular penalty and achieving high efficiency without the need to shift the switching frequency to very high values. The main disadvantage of shifting Fsw to the range of MHz is that all of the parasitic components (which are difficult to manage in production) have more and more impact on the functionality and efficiency to the power stage.

On the other side, the second area where fS < fR add the challenge to have enough gain to allow the LLC converter covering low VIN and high VOUT. High gain of the power stage can be obtained by designing properly the transformer with appropriate value of Ln (Lm / Lr) because this defines the maximum attainable gain. If the integrated leakage inductance of the transformer is not sufficient to get the right Ln value, then it is possible to add an extra inductor.

Figure 4-2 LLC Gain In and Out of Resonance
Equation 6. L   ×   I L 2 > C T O T A L ×   V S W 2 = 2 C O S S ×   V S W 2
Low Turn-Off Stress Figure 4-3 Low Turn-Off Stress
  • Necessary and sufficient soft-switching conditions:
    • Inductive input impedance
    • Sufficient energy inside resonant tank for discharging or charging output capacitor of switches in switching network
    • Enough dead-time between switches inside switch network

By leveraging TI GaN, we can achieve:

Optimizing internal GaN architecture by integrating drivers:

  • Improves reliability by reducing overshoot and ringing on drain-source voltage

Taking advantage of low Input and Output Capacitance:

  • Reduces switching losses in hard-switched converters
  • Allows faster switching frequency in hard-switched and soft-switched converters
  • Reduces circulating currents in soft-switched converters

Zero Reverse Recovery Charge:

  • No reverse-recovery losses in hard-switched, half-bridge converters
  • Enables new bridge-oriented topologies

Greatly-reduced switching loss:

  • Lower gate-drain capacitance (COSS) reduces transition period
  • Allows faster switching speeds
  • Reduced or eliminated heat sinking
Block Diagram of the Complete Solution Figure 4-4 Block Diagram of the Complete Solution

The power conversion architecture employs a multi-stage approach, engineered for high efficiency and compact form factors.

The two-stage EMI filter is composed of two stages, employing two common mode chokes and two x-capacitors. This is followed by a self-driven semi-active bridge rectifier, which almost eliminate the conduction losses on the two low side diodes of the bridge, but the high side diodes remain standard diodes. This method reduces the total loss in the bridge, and allows the use of only SMD components, avoiding expensive and bulky heat sinks.

The main PFC Boost stage, driven by UCC28056 controller is leveraging GaN to achieve enhanced efficiency by keeping high switching frequency and facilitating the use of a small inductor.

The second conversion stage is the LLC resonant converter, driven by the latest controller UCC256611.

A synchronous rectifier section is added to improve efficiency and keep losses and dissipation under control.

The system control and communication are handled by a MSPM0 microcontroller, sending VOUT and IOUT reference set points by means of I2C communication to two DACs.

Finally, an extra hot-swap function has been added to limit the worst case load current and managing load connections dynamically. This feature is crucial for robust operation in scenarios involving real battery connections or when a battery management system (BMS) oversees charging and discharging parameters.