SBAS661C February 2015 – May 2021 ADS1262 , ADS1263
PRODUCTION DATA
The key considerations In the design of a 3-wire RTD circuit are the accuracy, the lead wire compensation, and the sensor self-heating. As the design values of Table 10-2 illustrate, several values of excitation currents are available. The resolution is expressed in units of noise-free bits (NFR). Noise-free resolution is resolution with no code flicker. The selection of excitation currents trades off resolution against sensor self-heating. In general, measurement resolution improves with increasing excitation current. Increasing the excitation current beyond 1000 µA results in no further improvement in resolution. The design procedure is based on 500-µA excitation current, because this level of current results in very low sensor self-heating (0.4 mW).
I_{IDAC} (µA) | NFR (bits) | P_{RTD} (mW) | V_{RTD}^{(1)} (V) | Gain^{(2)} (V/V) | V_{REFMIN}^{(3)} (V) | V_{REF}^{(4)} (V) | R_{REF}^{(5)} (kΩ) | V_{INNLIM}^{(6)} (V) | V_{INPLIM}^{(7)} (V) | R_{BIAS}^{(8)} (kΩ) | V_{RTDN}^{(9)} (V) | V_{RTDP}^{(10)} (V) | V_{IDAC1}^{(11)} (V) |
---|---|---|---|---|---|---|---|---|---|---|---|---|---|
50 | 16.8 | 0.001 | 0.02 | 32 | 0.64 | 0.90 | 18 | 0.6 | 4.1 | 7.10 | 0.7 | 0.7 | 1.9 |
100 | 17.8 | 0.004 | 0.04 | 32 | 1.28 | 1.41 | 14.1 | 0.9 | 3.8 | 5.10 | 1.0 | 1.1 | 2.8 |
250 | 18.8 | 0.025 | 0.10 | 16 | 1.60 | 1.76 | 7.04 | 1.1 | 3.7 | 2.30 | 1.2 | 1.3 | 3.3 |
500 | 19.1 | 0.100 | 0.20 | 8 | 1.60 | 1.76 | 3.52 | 1.0 | 3.8 | 1.10 | 1.1 | 1.3 | 3.4 |
750 | 18.9 | 0.225 | 0.30 | 4 | 1.20 | 1.32 | 1.76 | 0.8 | 4.0 | 0.57 | 0.9 | 1.2 | 2.8 |
1000 | 19.3 | 0.400 | 0.40 | 4 | 1.60 | 1.76 | 1.76 | 0.9 | 3.9 | 0.50 | 1.0 | 1.4 | 3.5 |
1500 | 19.1 | 0.900 | 0.60 | 2 | 1.20 | 1.32 | 0.88 | 0.6 | 4.2 | 0.23 | 0.7 | 1.3 | 3.0 |
2000 | 18.3 | 1.600 | 0.80 | 1 | 0.80 | 0..90 | 0.45 | 0.3 | 4.5 | 0.10 | 0.4 | 1.2 | 2.4 |
Initially, R_{LEAD1} and R_{LEAD2} are considered to be 0 Ω. Route the IDAC1 current through the external reference resistor, R_{REF}. IDAC1 generates the ADC reference voltage, V_{REF}, across the reference resistor. This voltage is defined by Equation 24:
Route the second current (IDAC2) to the second RTD lead.
Program both IDAC1 and IDAC2 to the same value by using the IDACMAG register; however, only the IDAC1 current flows through the reference resistor and RTD. The IDAC1 current excites the RTD to produce a voltage proportional to the RTD resistance. The RTD voltage is defined by Equation 25:
The ADC amplifies the RTD signal voltage (V_{RTD}) and measures the resulting voltage against the reference voltage to produce a proportional digital output code, as shown in Equation 26 through Equation 28.
As shown in Equation 28, the RTD measurement depends on the value of the RTD, the PGA gain, and the reference resistor R_{REF}, but not on the IDAC1 value. Therefore, the absolute accuracy and temperature drift of the excitation current does not matter.
The second excitation current (IDAC2) provides a second voltage drop across the second RTD lead resistance, R_{LEAD2}. The second voltage drop compensates the voltage drop caused by I_{DAC1} and R_{LEAD1}. The leads of a 3-wire RTD typically have the same length; therefore, the lead resistance is typically identical. Taking the lead resistance into account (R_{LEADx} ≠ 0), the differential voltage (V_{IN}) across ADC inputs AIN4 and AIN5 is shown in Equation 29:
If R_{LEAD1} = R_{LEAD2} and I_{IDAC1} = I_{IDAC2}, the expression for V_{IN} reduces to Equation 30:
In other words, the measurement error resulting from the voltage drop across the RTD lead resistance is compensated, as long as the lead resistance values and the IDAC values are matched.
Using Equation 25, the value of RTD resistance (400 Ω, maximum) and the excitation current (500 μA) yields an RTD voltage of V_{RTD} = 500 μA · 400 Ω = 0.2 V. Use the maximum gain of 8 V/V in order to limit the reference voltage requirement as well as the corresponding loop voltage of IDAC1. The total loop voltage must not exceed the maximum IDAC voltage compliance specification. Gain = 8 requires a minimum reference voltage V_{REFMIN} = 0.2 V · 8 = 1.6 V. To provide a margin for the ADC operating range, increase the target reference voltage by 10% (V_{REF} = 1.6 V · 1.1 = 1.76 V). Calculate the value of the reference resistor, as shown in Equation 31:
For best results, use a precision reference resistor R_{REF} with a low temperature drift (< 10 ppm/°C).
The next step in the design is determining the value of the R_{BIAS} resistor, in order to level shift the RTD voltage to meet the ADC absolute input-voltage specification. The required level-shift voltage is determined by calculating the minimum absolute voltage (V_{INNLIM}) as shown in Equation 32:
where
The result of the equation requires a minimum absolute input voltage (V_{RTDN}) > 1.0 V. Therefore, the RTD voltage must be level shifted a minimum of 1.0 V. To meet this requirement, a target level-shift value of 1.1 V is chosen to provide 0.1 V margin. Calculate the value of R_{BIAS} as shown in Equation 33:
After the level-shift voltage is determined, verify that the positive RTD voltage (V_{RTDP}) is less than the maximum absolute input voltage (V_{INPLIM}), as shown in Equation 34:
where
Solving Equation 34 results in a required V_{RTDP} of less than 3.8 V. Calculate the V_{RTDP} input voltage by Equation 35:
Because 1.3 V is less than the 3.8-V maximum input voltage limit, the absolute positive and negative RTD voltages are within the ADC specified input range.
The next step in the design is to verify that the loop voltage of the excitation current is less than the specified IDAC compliance voltage. The IDAC compliance voltage is the maximum voltage drop developed across each IDAC current path to AVSS. In this circuit, IDAC1 has the largest voltage drop developed across its current path. The IDAC1 calculation is sufficient to satisfy IDAC2 because the IDAC2 voltage drop is always less than IDAC1 voltage drop. The sum of voltages in the IDAC1 loop is shown in Equation 36:
where
The equation results in a loop voltage of V_{IDAC1}= 3.4 V. The worst-case current source compliance voltage is: (V_{AVDD} – 1.1 V) = (4.75 V – 1.1 V) = 3.64 V. The V_{IDAC1} loop voltage is less than the specified current source compliance voltage (3.4 V < 3.64 V).
Many applications benefit from using an analog filter at the inputs to remove noise and interference from the signal. Filter components are placed on the ADC inputs (R_{F1}, R_{F2}, C_{DIF1}, C_{CM1}, and C_{CM2}), as well as on the reference inputs (R_{F3}, R_{F4}, C_{DIF2}, C_{CM3}, and C_{CM4}). The filters remove both differential and common-mode noise. The application shows a differential input noise filter formed by R_{F1}, R_{F2} and C_{DIF}, with additional differential mode capacitance provided by the common-mode filter capacitors, C_{M1} and C_{M2}. Calculate the differential cutoff frequency as shown in Equation 37:
The common-mode noise filter is formed by components R_{F1}, R_{F2}, C_{M1} and C_{M2}. Calculate the common-mode signal cutoff frequency as shown in Equation 38:
Mismatches in the common-mode filter components convert common-mode noise into differential noise. To reduce the effect of mismatch, use a differential mode filter with a corner frequency that is 10 times lower than the common-mode filter corner frequency. The low-frequency differential filter removes the common-mode converted noise. The filter resistors (R_{Fx}) also serve as current-limiting resistors. These resistors limit the current into the analog inputs (AINx) of the device to safe levels when an overvoltage occurs on the inputs.
Filter resistors lead to an offset voltage error due to the dc input current leakage flowing into and out of the device. Remove this voltage error by system offset calibration. Resistor values that are too large generate excess thermal noise and degrade the overall noise performance. The recommended range of the filter resistor values is 2 kΩ to 10 kΩ. The properties of the capacitors are important because the capacitors are connected to the signal; use high-quality C0G ceramics or film-type capacitors.
For consistent noise performance across the full range of RTD measurements, match the corner frequencies of the input and reference filter. Detailed information on matching the input and reference filter is found in the RTD Ratiometric Measurements and Filtering Using the ADS1148 and ADS1248 application report.